High order miller n-path filter

ABSTRACT

An N-path filter with one or more branches selectively coupled to a shared circuit node includes a first branch having a first feedback path and a second feedback path. The first feedback path includes a Miller amplifier having an input coupled to an input voltage and a first capacitor coupled to both the input voltage and an output of the Miller amplifier. The second feedback path includes a node in common with the first feedback path. The second feedback path also includes a first high pass filter coupled to the output of the Miller amplifier and a second capacitor coupled to both the first capacitor and the first high pass filter.

TECHNICAL FIELD

The present disclosure generally relates to wireless receivers. Morespecifically, the present disclosure relates to an application of Millereffect on N-path filters configured as bandpass filters to achieve ahigh order Miller N-path filter.

BACKGROUND

Advances in technology have resulted in smaller and more powerfulcomputing devices. For example, there currently exist a variety ofportable personal computing devices or user equipments (UEs), includingwireless computing devices, such as portable wireless telephones,personal digital assistants (PDAs), and paging devices that are small,lightweight, and easily carried by users. More specifically, portablewireless telephones, such as cellular telephones and Internet protocol(IP) telephones, can communicate voice and data packets over wirelessnetworks.

In some types of wireless networks, the UE communicates with one or morenetwork base stations. In some scenarios, different base stations mayuse different radio access technologies (RATs). The term RAT refers tothe physical connection for a radio-based communication network.Examples of different RATs include, without limitation, third generationpartnership project (3GPP) technologies (e.g., third generationtechnology (3G), fourth generation technology (4G), and fifth generationtechnology (5G)), millimeter wave (mmW) technology (extremely highfrequency (EHF)), Bluetooth technology, and Wi-Fi technology. In amillimeter wave (mmW) system, multiple antennas are used for beamforming(e.g., in the range of 30 gigahertz (GHz), 60 GHz, etc.)

The different RATs may have different capabilities. For example, the UEmay have the capability to access both a long term evolution (LTE)network and a millimeter wavelength (mmW) network. The downlink/uplink(DL/UL) access link between an LTE base station and the UE is generallymore reliable than the access link between an mmW base station and theUE. However, the LTE link generally has lower capacity than the mmWlink.

In UEs that can simultaneously transmit and receive wirelesscommunications in accordance with the wireless network, transmit (TX)leakage can impose a performance limitation on receive (RX) circuitry.TX leakage and other jammers in the RX circuitry can be modulated anddown-converted to baseband along with a received wireless signal. TXleakage and jammers may have a relatively large voltage swing comparedto the received signal and may saturate an output of a receiver thatconverts the received signal from radio frequency (RF) to baseband.

In carrier aggregation (CA) architectures, blockers (TX leakage andjammers) are a performance limitation of RX circuitry. For intra-CAoperation where a low noise amplifier (LNA) is followed by a cascodedevice or a transconductance stage, the LNA output is a high impedancenode. The high impedance causes large blocker swing and linearityissues. For concurrent CA operation, the noise figure in one CA receivepath may be degraded if a signal in another CA receive path is largerand acts as a jammer.

SUMMARY

An N-path filter with one or more branches selectively coupled to ashared circuit node includes a first branch having a first feedback pathand a second feedback path. The first feedback path includes anamplifier (e.g., a Miller amplifier or an amplifier based on Millereffect) having an input coupled to an input voltage and a firstcapacitor coupled to both the input voltage and an output of the Milleramplifier. The second feedback path includes a node in common with thefirst feedback path. The second feedback path also includes a first highpass filter coupled to the output of the Miller amplifier and a secondcapacitor coupled to both the first capacitor and the first high passfilter.

An N-path filter with one or more branches selectively coupled to ashared circuit node includes a first branch having a first feedback pathand a second feedback path. The first feedback path includes a Milleramplifier having an input coupled to an input voltage and first meansfor generating an impedance coupled to both the input voltage and anoutput of the Miller amplifier. The second feedback path includes a nodein common with the first feedback path. The second feedback path alsoincludes a first high pass filter coupled to the output of the Milleramplifier and second means for generating an impedance coupled to boththe first impedance generating means and the first high pass filter.

A method for filtering a wireless signal at a receiver includesreceiving a radio frequency signal at a shared circuit node selectivelycoupled to each of multiple branches of an N-path filter. Each branch ofthe N-path filter includes a first feedback path including a Milleramplifier and a second feedback path having a node in common with thefirst feedback path. The method also includes generating a highimpedance at the common node to prevent the radio frequency signal fromtraversing the first feedback path and the second feedback path when theradio frequency signal is received at a first frequency. The methodfurther includes generating a low impedance at the common node to allowthe radio frequency signal to traverse the first feedback path and/orthe second feedback path when the radio frequency signal is received ata second frequency. The second frequency is higher than the firstfrequency.

This has outlined, rather broadly, the features and technical advantagesof the present disclosure in order that the detailed description thatfollows may be better understood. Additional features and advantages ofthe disclosure will be described below. It should be appreciated bythose skilled in the art that this disclosure may be readily utilized asa basis for modifying or designing other structures for carrying out thesame purposes of the present disclosure. It should also be realized bythose skilled in the art that such equivalent constructions do notdepart from the teachings of the disclosure as set forth in the appendedclaims. The novel features, which are believed to be characteristic ofthe disclosure, both as to its organization and method of operation,together with further objects and advantages, will be better understoodfrom the following description when considered in connection with theaccompanying figures. It is to be expressly understood, however, thateach of the figures is provided for the purpose of illustration anddescription only and is not intended as a definition of the limits ofthe present disclosure.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the present disclosure, referenceis now made to the following description taken in conjunction with theaccompanying drawings.

FIG. 1 shows a wireless device communicating with a wirelesscommunication system.

FIG. 2 shows a block diagram of the wireless device in FIG. 1, accordingto an aspect of the present disclosure.

FIGS. 3A-3D show four examples of carrier aggregation (CA).

FIG. 4A illustrates an example receive chain of a radio frequency (RF)front end including an N-path filter, in accordance with certain aspectsof the present disclosure.

FIG. 4B is an example timing diagram illustrating signals used tocontrol switching operations of transistors in the N-path filter of FIG.4A, in accordance with certain aspects of the present disclosure.

FIG. 5A illustrates an example frequency response of an N-path filter,in accordance with certain aspects of the present disclosure.

FIG. 5B illustrates an example frequency response of an N-path filteraccording to aspects of the present disclosure.

FIG. 6 illustrates an N-path filter including a single high passfiltering path according to aspects of the present disclosure.

FIG. 7 illustrates an N-path filter including two high pass filteringpaths according to aspects of the present disclosure.

FIG. 8 is a graph showing a first frequency response of a lower orderbandpass filter and a second frequency response of a higher orderbandpass filter.

FIG. 9 represents a Thévenin equivalent circuit of the N-path filter ofFIG. 7 including two high pass filtering paths according to aspects ofthe present disclosure.

FIG. 10 illustrates an N-path filter with multiple branches, with eachbranch including two high pass filtering paths.

FIG. 11 illustrates an N-path filter with multiple branches, with eachbranch including two high pass filtering paths.

FIG. 12 depicts a simplified flowchart of a method for filtering jammersignals according to aspects of the present disclosure.

FIG. 13 is a block diagram showing an exemplary wireless communicationsystem in which a configuration of the disclosure may be advantageouslyemployed.

DETAILED DESCRIPTION

The detailed description set forth below, in connection with theappended drawings, is intended as a description of variousconfigurations and is not intended to represent the only configurationsin which the concepts described herein may be practiced. The detaileddescription includes specific details for the purpose of providing athorough understanding of the various concepts. However, it will beapparent to those skilled in the art that these concepts may bepracticed without these specific details. In some instances, well-knownstructures and components are shown in block diagram form in order toavoid obscuring such concepts. As described herein, the use of the term“and/or” is intended to represent an “inclusive OR”, and the use of theterm “or” is intended to represent an “exclusive OR”.

In user equipments (UEs) that can simultaneously transmit and receivewireless signals in accordance with the wireless network, transmit (TX)leakage can impose a performance limitation on receive (RX) circuitry.TX leakage and other jammers (or jammer signals) in the RX circuitry canbe modulated and down-converted to baseband along with a receivedwireless signal. TX leakage and jammers may have a relatively largevoltage swing compared to the received signal and may saturate an outputof a receiver that converts the received signal from radio frequency(RF) to baseband.

In carrier aggregation (CA) architectures, blockers (transmit (TX)leakage and jammers) are a performance limitation of RX circuitry. TXleakage is the leakage from a TX chain (e.g., a transmit path includinga transmitter) into a receive (RX) chain (e.g., a receive path includinga receiver) in a transceiver or transceiver front end. Signals at alocal oscillator (LO) frequency and multiples thereof (e.g., two andthree times the LO frequency) coupling into the RX chain together withthe TX leakage can be modulated and down-converted to the baseband (BB).Large voltage swings can saturate the BB output. This may be mitigatedby decreasing the BB gain, although this may degrade the signal-to-noiseratio (SNR).

In non-CA applications, TX leakage or blockers and jammers are limitingfactors to improving noise figure (NF) and linearity. Large blockers andTX leakage may prevent employing some of the architectures that arewell-suited to achieving the increased NF. For example, the TX jammerscause TX second order intercept point (IIP2) problems and limitssignal-to-noise distortion ratio (SNDR) performance of a receiver.

Thus, circuits used to reject TX leakage or blockers to maintainsensitivity during concurrent CA or non-CA operation are important. Sometechniques for reducing the effect of the jammer include a large TXcapacitor used in front of a transimpedance amplifier (TIA) to absorb aTX jammer current. This method, however, specifies a large TX capacitorto sink a current of the transmit jammer into the TX capacitor to reducea voltage swing (as the voltage swing is inversely proportional to thecapacitance of the TX capacitor). In addition, if the TX jammer is tooclose to in-band, the amount of the TX rejection is small, which causesa lower gain mode to be specified, degrading noise figure (NF). Thissmall amount of TX rejection holds true even when the large TX capacitoris used. Therefore, an optimum signal-to-noise ratio cannot be achievedwith the circuit.

Moreover, the bandwidth of desired signals is becoming larger and larger(e.g., on the order of 80 MHz). Therefore, it is becoming increasinglychallenging to implement a high quality factor (high-Q) bandpass filterat radio frequencies to reject out-of-band jammers and TX leakage whileavoiding attenuation of desired signals in the RX band.

N-path filters may be used to provide high-Q bandpass filters at radiofrequencies. An N-path filter may include N-branches. For example, theN-path filter may be composed of N identical linear time invariant (LTI)networks and 2N frequency mixers driven by time/phase-shifted versionsof a clock signal. If the LTI networks exhibit a low-pass characteristicaround direct current (DC), mixing by the mixers results in a bandpassfilter response with a passband centered around the mixing frequency.That is, the input signal is down-converted to baseband, filtered by theLTI network, and then up-converted again to the original band of theinput signal. The center frequency is determined by the mixing frequencyand is insensitive to filter component values. A high mixing frequencycombined with a narrow low-pass filter bandwidth provides a very high-Qfilter.

Some techniques use Miller effect to reduce the size of the capacitorused in an N-path filter to improve performance of the RX circuitry. Forexample, amplification (using a Miller amplifier) may be applied toincrease a voltage swing across the capacitor. Accordingly, the size ofthe capacitor used for the N-path filter is reduced. However, thesetechniques may not achieve high-order filtering while applying theMiller effect to the capacitor. It is therefore desirable to design ahigh-order high-Q bandpass filter to mitigate these issues.

Aspects of the present disclosure include an N-path filter with one ormore branches selectively coupled to a shared circuit node. The N-pathfilter is coupled to a low noise amplifier (LNA). For example, theN-path filter may be coupled to an input of the LNA, an output of theLNA or between two LNAs of the receiver. The N-path filter may beconfigured as a bandpass filter. Each branch includes a first feedbackpath and a second feedback path (or high pass filtering path). When theone or more branches include multiple branches, each of the multiplebranches may be selectively coupled with a shared circuit node. Asynthesized impedance may be achieved by coupling the multiple branchesin parallel.

In one aspect of the present disclosure, the first feedback path of thefirst branch of the N-path filter includes an amplifier (e.g., a Milleramplifier or an amplifier based on Miller effect) and a first capacitor.The Miller amplifier includes a first terminal (e.g., an input) coupledto an input voltage. The first terminal may correspond to the sharedcircuit node. For example, the N-path filter may be connected or coupledto a circuit (e.g., a Thévenin equivalent circuit) that includes aninput signal from a voltage source Vin through an input impedance (e.g.,a series resistance). The Thévenin equivalent circuit may represent anequivalent of a signal received by an antenna and amplified by the LNA.The voltage source Vin may be deemed the input voltage.

The first capacitor is coupled to both the input voltage and a secondterminal (e.g., an output) of the Miller amplifier. For example, theinput of the Miller amplifier is coupled to the shared circuit node,which is coupled to the voltage source or input voltage Vin. In oneaspect of the disclosure, the first capacitor is coupled to the inputvoltage through a first switch or transistor and coupled to the outputof the Miller amplifier through a second transistor.

The second feedback path has a first node in common with the firstfeedback path. The second feedback path includes a first high passfilter and a second capacitor. The first node in common with the firstfeedback path merges the second feedback path to the first feedbackpath. For example, the second capacitor is coupled to the input of theMiller amplifier via the first transistor. The first high pass filter iscoupled to the output of the Miller amplifier. For example, the firsthigh pass filter may be coupled to the output of the Miller amplifiervia the second transistor.

In one aspect of the disclosure, the second feedback path may furtherinclude a first buffer. The first buffer may be positioned between thefirst high pass filter and the second capacitor. The first buffer may beimplemented in accordance with a biased source follower configuration.

In some aspects of the disclosure, the N-path filter may include a thirdfeedback path and a third capacitor. The third feedback path may bemerged with the second feedback path. The third feedback path includes asecond high pass filter coupled to an output of the first high passfilter. The third capacitor may be coupled to both the first capacitorand the second capacitor as well as the second high pass filter. Thefirst buffer is also coupled between an output of the first high passfilter and an input of the second high pass filter. The N-path filtermay be integrated into a receive (RX) chain before or after a low noiseamplifier (LNA) in accordance with a shunt configuration.

The aspects of the present disclosure may be implemented in the systemof FIG. 1 and the system of FIG. 13. More specifically, aspects of thepresent disclosure may be implemented in the wireless device 200 of FIG.2.

FIG. 1 shows a wireless device 110, which may include the disclosedN-path filter, communicating with a wireless communication system 120.The wireless communication system 120 may be a 5G system, a long termevolution (LTE) system, a code division multiple access (CDMA) system, aglobal system for mobile communications (GSM) system, a wireless localarea network (WLAN) system, or some other wireless system. A CDMA systemmay implement wideband CDMA (WCDMA), time division synchronous CDMA(TD-SCDMA), CDMA2000, or some other version of CDMA. For simplicity,FIG. 1 shows the wireless communication system 120 including two basestations 130 and 132 and one system controller 140. In general, awireless system may include any number of base stations and any numberof network entities.

A wireless device 110 may be referred to as a user equipment (UE), amobile station, a terminal, an access terminal, a subscriber unit, astation, etc. The wireless device 110 may also be a cellular phone, asmartphone, a tablet, a wireless modem, a personal digital assistant(PDA), a handheld device, a laptop computer, a Smartbook, a netbook, acordless phone, a wireless local loop (WLL) station, a Bluetooth device,etc. The wireless device 110 may be capable of communicating with thewireless communication system 120. The wireless device 110 may also becapable of receiving signals from broadcast stations (e.g., a broadcaststation 134), signals from satellites (e.g., a satellite 150) in one ormore global navigation satellite systems (GNSS), etc. The wirelessdevice 110 may support one or more radio technologies for wirelesscommunication such as 5G, LTE, CDMA2000, WCDMA, TD-SCDMA, GSM, 802.11,etc.

The wireless device 110 may support carrier aggregation, which isoperation on multiple carriers. Carrier aggregation may also be referredto as multi-carrier operation. According to an aspect of the presentdisclosure, the wireless device 110 may be able to operate in low-bandfrom 698 to 960 megahertz (MHz), mid-band from 1475 to 2170 MHz, and/orhigh-band from 2300 to 2690 MHz, ultra-high band from 3400 to 3800 MHz,and long term evolution (LTE) in LTE unlicensed bands (LTE-U/LAA) from5150 MHz to 5950 MHz. Low-band, mid-band, high-band, ultra-high band,and LTE-U refer to five groups of bands (or band groups), with each bandgroup including a number of frequency bands (or simply, “bands”). Forexample, in some systems each band may cover up to 200 MHz and mayinclude one or more carriers. For example, each carrier may cover up to40 MHz in LTE. Of course, the range for each of the bands is merelyexemplary and not limiting, and other frequency ranges may be used. LTERelease 11 supports 35 bands, which are referred to as LTE/UMTS bandsand are listed in 3GPP TS 36.101. The wireless device 110 may beconfigured with up to 5 carriers in one or two bands in LTE Release 11.

FIG. 2 shows a block diagram of an exemplary design of a wireless device200, such as the wireless device 110 shown in FIG. 1. FIG. 2 shows anexample of a transceiver 220, which may be a wireless transceiver (WTR).In general, the conditioning of the signals in a transmitter 230 and areceiver 250 may be performed by one or more stages of amplifier(s),filter(s), up-converters, down-converters, and the like. These circuitblocks may be arranged differently from the configuration shown in FIG.2. Furthermore, other circuit blocks not shown in FIG. 2 may also beused to condition the signals in the transmitter 230 and receiver 250.Unless otherwise noted, any signal in FIG. 2, or any other illustrationsin the drawings, may be either single-ended or differential. Somecircuit blocks in FIG. 2 may also be omitted.

In the example shown in FIG. 2, the wireless device 200 generallyincludes the transceiver 220 and a data processor 210. The dataprocessor 210 may include a memory (not shown) to store data and programcodes, and may generally include analog and digital processing elements.The transceiver 220 may include the transmitter 230 and receiver 250that support bi-directional communication. In general, the wirelessdevice 200 may include any number of transmitters and/or receivers forany number of communication systems and frequency bands. All or aportion of the transceiver 220 may be implemented on one or more analogintegrated circuits (ICs), radio frequency (RF) integrated circuits(RFICs), mixed-signal ICs, and the like.

A transmitter or a receiver may be implemented with a super-heterodynearchitecture or a direct-conversion architecture. In thesuper-heterodyne architecture, a signal is frequency-converted betweenradio frequency and baseband in multiple stages, e.g., from radiofrequency to an intermediate frequency (IF) in one stage, and fromintermediate frequency to baseband in another stage for a receiver. Inthe direct-conversion architecture, a signal is frequency-convertedbetween radio frequency and baseband in one stage. The super-heterodyneand direct-conversion architectures may use different circuit blocksand/or have different requirements. In the example shown in FIG. 2, thetransmitter 230 and the receiver 250 are implemented with thedirect-conversion architecture.

In a transmit path, the data processor 210 processes data to betransmitted. The data processor 210 also provides in-phase (I) andquadrature (Q) analog output signals to the transmitter 230 in thetransmit path. In an exemplary aspect, the data processor 210 includesdigital-to-analog converters (DACs) 214 a and 214 b for convertingdigital signals generated by the data processor 210 into the in-phase(I) and quadrature (Q) analog output signals (e.g., I and Q outputcurrents) for further processing.

Within the transmitter 230, lowpass filters 232 a and 232 b filter thein-phase (I) and quadrature (Q) analog transmit signals, respectively,to reduce undesired images caused by the prior digital-to-analogconversion. Amplifiers (Amp) 234 a and 234 b amplify the signals fromlowpass filters 232 a and 232 b, respectively, and provide in-phase (I)and quadrature (Q) baseband signals. An up-converter 240 includingupconversion mixers 241 a and 241 b up-converts the in-phase (I) andquadrature (Q) baseband signals with in-phase (I) and quadrature (Q)transmit (TX) local oscillator (LO) signals from a TX LO signalgenerator 290 to provide an up-converted signal. A filter 242 filtersthe up-converted signal to reduce undesired images caused by thefrequency up-conversion as well as interference in a receive frequencyband. A power amplifier (PA) 244 amplifies the signal from filter 242 toobtain the desired output power level and provides a transmit radiofrequency signal. The transmit radio frequency signal is routed througha duplexer/switch 246 and transmitted via an antenna 248.

In a receive path, the antenna 248 receives communication or wirelesssignals and provides a received radio frequency (RF) signal, which isrouted through the duplexer/switch 246 and provided to a low noiseamplifier (LNA) 252. The duplexer/switch 246 is designed to operate witha specific receive (RX) to transmit (TX) (RX-to-TX) duplexer frequencyseparation, such that RX signals are isolated from TX signals. Thereceived RF signal is amplified by the LNA 252 and filtered by a filter254 to obtain a desired RF input signal. Down-conversion mixers 261 aand 261 b mix the output of the filter 254 with in-phase (I) andquadrature (Q) receive (RX) LO signals (i.e., LO_I and LO_Q) from an RXLO signal generator 280 to generate in-phase (I) and quadrature (Q)baseband signals. The in-phase (I) and quadrature (Q) baseband signalsare amplified by amplifiers 262 a and 262 b and further filtered bylowpass filters 264 a and 264 b to obtain in-phase (I) and quadrature(Q) analog input signals, which are provided to the data processor 210.In the exemplary configuration shown, the data processor 210 includesanalog-to-digital converters (ADCs) 216 a and 216 b for converting theanalog input signals into digital signals for further processing by thedata processor 210.

In FIG. 2, the transmit local oscillator (TX LO) signal generator 290generates the in-phase (I) and quadrature (Q) TX LO signals used forfrequency upconversion, while a receive local oscillator (RX LO) signalgenerator 280 generates the in-phase (I) and quadrature (Q) RX LOsignals used for frequency down-conversion. Each LO signal is a periodicsignal with a particular fundamental frequency. A phase locked loop(PLL) 292 receives timing information from the data processor 210 andgenerates a control signal used to adjust the frequency and/or phase ofthe TX LO signals from the TX LO signal generator 290. Similarly, a PLL282 receives timing information from the data processor 210 andgenerates a control signal used to adjust the frequency and/or phase ofthe RX LO signals from the RX LO signal generator 280.

The wireless device 200 may support carrier aggregation and may (i)receive multiple downlink signals transmitted by one or more cells onmultiple downlink carriers at different frequencies, and/or (ii)transmit multiple uplink signals to one or more cells on multiple uplinkcarriers. For intra-band carrier aggregation, the transmissions are senton different carriers in the same band. For inter-band carrieraggregation, the transmissions are sent on multiple carriers indifferent bands. Those skilled in the art will understand, however, thataspects described herein may be implemented in systems, devices, and/orarchitectures that do not support carrier aggregation.

The wireless device 300 may support carrier aggregation, which isoperation on multiple carriers. Carrier aggregation may also be referredto as multi-carrier operation.

In general, carrier aggregation (CA) may be categorized into two types:intra-band CA and inter-band CA. Intra-band CA refers to operation onmultiple carriers within the same band and inter-band CA refers tooperation on multiple carriers in different bands.

FIG. 3A shows an example of contiguous intra-band CA. In the exampleshown in FIG. 3A, a wireless device (e.g., the wireless device 110) isconfigured with four contiguous carriers in the same band, which is aband in low-band. The wireless device may send and/or receivetransmissions on multiple contiguous carriers within the same band.

FIG. 3B shows an example of non-contiguous intra-band CA. In the exampleshown in FIG. 3B, a wireless device (e.g., the wireless device 110) isconfigured with four non-contiguous carriers in the same band, which isa band in low-band. The carriers may be separated by 5 MHz, 10 MHz, orsome other amount. The wireless device may send and/or receivetransmissions on multiple non-contiguous carriers within the same band.

FIG. 3C shows an example of inter-band CA in the same band group. In theexample shown in FIG. 3C, a wireless device (e.g., the wireless device110) is configured with four carriers in two bands in the same bandgroup, which is low-band. The wireless device may send and/or receivetransmissions on multiple carriers in different bands in the same bandgroup (e.g., low-Band 1 (LB1) and low-Band 2 (LB2) in FIG. 3C).

FIG. 3D shows an example of inter-band CA in different band groups. Inthe example shown in FIG. 3D, a wireless device (e.g., the wirelessdevice 110) is configured with four carriers in two bands in differentband groups, which include two carriers in one band in low-band and twoadditional carriers in another band in high-band. The wireless devicemay send and/or receive transmissions on multiple carriers in differentbands in different band groups (e.g., low-band and high-band in FIG.3D). FIGS. 3A to 3D show four examples of carrier aggregation. Carrieraggregation may also be supported for other combinations of bands andband groups. For example, carrier aggregation may be supported forlow-band and high-band, mid-band and high-band, high-band and high-band,and other band combinations with ultra-high band and long term evolutionin unlicensed spectrum (LTE-U).

FIG. 4A illustrates an example receive chain (e.g., including thereceiver 250) of an RF front end comprising an N-path filter 402, inaccordance with certain aspects of the present disclosure. In certainaspects, the LNA 252 of FIG. 2 may include two separate LNAs (e.g., afirst LNA 452A and a second LNA 452B). For certain aspects, the firstLNA 452A may be external to an RF integrated circuit (RFIC), while thesecond LNA 452B may be included in the RFIC, along with other circuits(e.g., the mixers 261 a and 261 b as well as the lowpass filters 264 aand 264 b). The lowpass filters 264 a and 264 b may be baseband filters(BBF). For other aspects, the first LNA 452A and the second LNA 452B mayboth be included in the RFIC, along with the other circuits. The N-pathfilter 402 may be connected with a node 406 between the LNAs 452A and452B. In this manner, the N-path filter 402 may function as a shuntfilter having a frequency response 410 in an effort to pass signals in adesired receive (RX) band and reject signals having frequencies outsidethis band (including TX leakage and jammers).

The N-path filter 402 has a number N=4 of parallel branches selectivelyconnected with the node 406, which is a common node for the branches.Those having ordinary skill in the art of N-path filters understand thatthere may be more or less than N=4 branches in any of the variousaspects of the present disclosure.

The N-path filter 402 may include a number of switches 404 (e.g., Nswitches, one in each filter branch), which may be implemented withn-channel metal oxide semiconductor (NMOS) transistors, individuallylabeled as transistors M1, M2, M3, and M4 in FIG. 4A. For other aspects,the switches 404 in the N-path filter may be implemented with p-channelmetal-oxide-semiconductor (PMOS) transistors or a combination of PMOSand NMOS transistors.

The four transistors M1, M2, M3, and M4 may be controlled using four 25%duty cycle signals P1, P2, P3, and P4, respectively, as illustrated inthe timing diagram 412 of FIG. 4B. In this manner, one switch 404 may beopened before or as the next switch in the control signal sequence isclosed. That is, each of the transistors M1, M2, M3, and M4 may bedriven such that the transistors are activated in sequence and periodsduring which each transistor is activated (e.g., each switch 404 isclosed) ideally do not overlap, although a small amount of overlap maybe tolerated for practical implementations. The duty cycle of thecontrol signals may be a function of the number N of filter branches(e.g., equal to 1/N). The amount of overlap, if any, in the controlsignals P1, P2, P3, etc. may be a small fraction (e.g., 1/10th) of theduty cycle.

Each switch 404 may connect a corresponding impedance ZA, ZB, ZC, or ZDwith the node 406 when closed. Impedances ZA, ZB, ZC, and ZD may allhave the same impedance value. One end of each impedance ZA, ZB, ZC, orZD may be connected with a corresponding switch 404, and the other endof each impedance may be connected with a reference potential (e.g.,electrical ground, a power supply voltage, or a bias voltage) for theN-path filter 402.

In this configuration, the frequency response 410 of the N-path filter402 may have a center frequency approximately equal to the switchingfrequency of the control signals P1, P2, P3, and P4 for the transistorsM1, M2, M3, and M4, respectively. For example, the switching frequencymay be considered as the inverse of the period between rising edges ofthe control signal P1, shown by vertical dashed lines in timing diagram412. The control signals P1, P2, P3, and P4 may have the same frequency(e.g., the switching frequency), but different phases. Moreover, thebandwidth of the frequency response 410 may be twice the bandwidth of apole of the branch impedance (ZA ZB, ZC, or ZD).

FIG. 5A illustrates example frequency responses of N-path filters. Afrequency response 502 may correspond to an N-path filter configuredsuch that each of the branch impedances (e.g., ZA, ZB, ZC, and ZD) haveonly one pole (e.g., implemented with a resistor-capacitor (RC) load).An N-path filter with only one pole may provide a narrow-band, high-Qbandpass filter, but when used as a filter for a wide RX band 503, itmay not provide sufficient out-of-band rejection. Aspects of the presentdisclosure provide an N-path filter having a wideband frequency response504. For example, an N-path filter in accordance with the presentdisclosure may provide similar rejection at the TX leakage frequency(fTx) 506 as a narrow-band N-path filter, but with a wide, flat passbandshape.

FIG. 5B illustrates an example frequency response of an N-path filteraccording to aspects of the present disclosure. A frequency response 508may correspond to an N-path filter configured without a frequencydependent capacitor, a high pass filter, and/or a buffer. The frequencyresponse 508 includes transmit jammers that degrade the quality of thereceived radio frequency signal. A frequency response 510 corresponds toan N-path filter configured with a frequency dependent capacitor, a highpass filter, and/or a buffer. The frequency dependent capacitor, thehigh pass filter, and the buffer may be implemented in a feedback pathto provide a high pass feedback to create a sharper frequency responseor sharper filter. The sharper frequency response 510 eliminates thetransmit jammer.

Some N-path filters may only include switches and capacitors. Forexample, the N-path filter may only include switches such as M1 a and M1b (as shown in FIG. 6) and a capacitor C1 (as shown in FIG. 6). However,such N-path filters may not achieve flat frequency response or widerbandwidth in-band and high rejection of a transmit frequency or atransmit jammer. Accordingly, aspects of the present disclosure includeadditional circuitry to achieve flat frequency response or widerbandwidth in-band and high rejection of the transmit jammer

FIG. 6 illustrates an N-path filter 600 including a single high passfiltering path according to aspects of the present disclosure. Forillustrative purpose, only a single branch of the N-path filter 600 isshown. The N-path filter 600, however, may include multiple branchesoperating in accordance with different phases (e.g., zero degrees,ninety degrees, one hundred and eighty degrees, two hundred and seventydegrees, etc.) For example, the N-path filter 600 includes a singlebranch (at a phase of zero degrees) that may be used to provide animpedance (e.g., ZA, ZB, Zc, or ZD) of the N-path filter 600. Asillustrated, the N-path filter 600 includes a first transistor M1 a, asecond transistor M1 b, a third transistor M1 c, a fourth transistor M1d, a first capacitor C1, a second capacitor C2, a first high passcapacitor CHP1, and a first resistor RHP1.

The N-path filter 600 is illustrated as being connected with a Théveninequivalent circuit 602 (via the node 606) having an input signal (e.g.,voltage source Vin) and an input impedance (e.g., series resistance RS),which may represent the equivalent of the signal received by an antenna(e.g., the antenna 248) and amplified by an LNA (e.g., the first LNA452A). In some aspects, the node 606 may be between two LNAs of areceiver (e.g., between the LNAs 452A and 452B), prior to an LNA of thereceiver (e.g., prior to the LNA 452A) or after the LNA(s) of thereceiver (e.g., after the LNA 452B). Alternatively, the node 606 may becoupled to a mixer (e.g., the mixer 261 a or the mixer 261 b).

The first transistor M1 a, the second transistor M1 b, and the firstcapacitor C1 are connected in series with a Miller amplifier 620 to forma first feedback path. A second feedback path (or high pass filteringpath) of the first branch includes the third transistor M1 c, the fourthtransistor M1 d, the second capacitor C2, the first high pass capacitorCHP1, and the first resistor RHP1. The first high pass capacitor CHP1and the first resistor RHP1 form a high pass filter. In this case, thehigh pass filtering path includes only one zero (e.g., a zero at 5.3MHz). The third transistor M1 c and the fourth transistor M1 d form abuffer coupled to the high pass filter. In one aspect of the disclosure,the buffer is a source follower (e.g., a voltage buffer or biased sourcefollower device). The source follower isolates the second capacitor C2from the first high capacitor CHP1 and the first resistor RHP1.Otherwise, the second capacitor C2 may alter the high-pass filtercharacteristics, which makes it difficult to program the zero frequency.

The second feedback path is merged with the first feedback path suchthat a current at a common node (e.g., node(s) 622 and/or 624) is sharedbetween the first feedback path and the second feedback path. Forexample, the second feedback path is coupled to the first feedback pathvia nodes 622 and 624. The node 622 is at a source (or drain) of thefirst transistor M1 a and the node 624 is at a drain (or source) of thesecond transistor M1 b. The second capacitor C2 is coupled between thenode 622 and a node 626 while the first capacitor C1 is coupled betweenthe node 622 and the node 624. The first high pass capacitor CHP1 isbetween a node 616 and the node 624. The node 606 is coupled to thefirst terminal (e.g., an input) of the Miller amplifier 620 and to adrain (or source) of the first transistor M1 a. A source (or drain) ofthe second transistor M1 b is coupled to a second terminal (e.g., anoutput) of the Miller amplifier 620.

The voltage at node 624 passes through the high-pass filter created bythe first resistor RHP1 and the first high pass capacitor CHP1. Thethird transistor M1 c senses a high-pass signal voltage at the node 616and the source follower provides a same voltage at the node 626, whichdrives the second capacitor C2. The first capacitor C1 and the secondcapacitor C2 are coupled to node 622, which means that current flowingthrough the first capacitor C1 and the second capacitor C2 converges atnode 622. A voltage at node 626 is a high pass version of a voltage atnode 624, which is a voltage after the Miller amplifier. Thus, thecurrent flowing through the second capacitor C2 is different from thecurrent flowing through the first capacitor C1. For example, the currentthrough the second capacitor C2 only includes current withhigh-frequency components. As a result, more high frequency componentsare present at node 622 through which the second capacitor C2 and thecorresponding high path filtering path are coupled, achieving strongerfeedback at high frequency. Thus, low impedance at a transmission (TX)jammer frequency results, which leads to a low TX jammer voltage swingat nodes 606 and 622.

A source of the third transistor M1 c is coupled to a drain of thefourth transistor M1 d at node 626. A power supply 612 is coupled to adrain of the third transistor M1 c while a source of the fourthtransistor M1 d is coupled to ground 614 (or a reference potential). Thebuffer is coupled to the high pass filter via a gate of the thirdtransistor M1 c. For example, the gate of the third transistor M1 c iscoupled to the node 616 between the first high pass capacitor CHP1 andthe first resistor RHP1. The third transistor M1 c is biased based on abias voltage VB provided at a node 618 of the high pass filter. The highpass filter and buffer are configured such that a frequency dependentcapacitor (e.g., the second capacitor C2) can be implemented in theN-path filter 600. As the frequency increases, the impedance of thesecond capacitor C2 decreases, which cause sharper filtering.

The first transistor M1 a and the second transistor M1 b may operate asa mixer to down-convert or up-convert a signal received by the N-pathfilter 600. To operate as a mixer, a gate of each of the firsttransistor M1 a and the second transistor M1 b are coupled to a same LO(or different LOs) to operate at a same or different LO frequency. Thefirst transistor M1 a and the second transistor M1 b may be driven by asame local oscillator (LO) phase and frequency. For example, the N-pathfilter 600 may be operating at a phase of zero (0) degrees, such thatthe first transistor M1 a and the second transistor M1 b are in phase(e.g., at a phase of zero (0) degrees) with each other. However, thefirst transistor M1 a and the second transistor M1 b may be driven bydifferent LO frequencies. For example, the first transistor M1 a and thesecond transistor M1 b may be driven by different LO frequencies tocause the transistors to operate at different frequencies to achievedown-conversion or upconversion of a radio frequency or baseband signalreceived by the N-path filter. Whether the upconversion occurs in thefirst transistor M1 a or the second transistor M1 b depends on thesignal received at the node 606.

For example, nodes 622 and 624 may include or may be characterized bybaseband voltages, while nodes 606 and 628 include or are characterizedby radio frequency signals. At an in-band frequency that is close to DC,the impedance associated with the first capacitor C1 and the secondcapacitor C2 is large. In this case, a received signal (which is closeto DC) cannot pass through the first capacitor C1 and the secondcapacitor C2 and therefore there is little or no feedback (e.g., thehigh impedance acts like an open circuit). Accordingly, the firsttransistor M1 a down-converts the signal received at the node 606 (localoscillator frequency plus a low frequency which is close to DC) suchthat the signal at the node 622 is a down-converted version of thesignal at the node 606. However, in the presence of an out of bandfrequency (e.g., transmitter (TX) jammer frequency including a localoscillator frequency plus a high frequency (e.g., 100 MHz)), theimpedances of the first capacitor C1 and the second capacitor C2 aresmall and thus the feedback gets stronger. In this case, the signalflows through node 628 to node 624 (down-conversion), to the firstcapacitor C1 and second capacitor C2, to node 622 and then to node 606(upconversion). The signal at node 606 feeds the Miller amplifier(feedback loop).

Aspects of the present disclosure use Miller effect to achieve highimpedance (e.g., capacitance) with a reduced impedance (e.g., a reducedsized capacitor) rather than a large impedance (e.g., a largecapacitor). The Miller effect accounts for an increase in an equivalentinput impedance (e.g., capacitance) of an inverting voltage amplifierdue to amplification of the effect of impedance between the input andoutput terminals. For example, a Miller amplifier is configured toamplify a Miller capacitance formed at an input node of the Milleramplifier.

The Miller amplifier 620 is included in the N-path filter 600 to reducethe filter bandwidth without increasing the value of the capacitors. TheMiller amplifier 620 increases loop gain associated with the N-pathfilter 600. In one aspect, the Miller amplifier applies amplification(e.g., gain of −4) to create more voltage swing across the firstcapacitor C1 and/or across the second capacitor C2. For example, thecapacitors (C1 and/or C2) experience one volt (positive) and four volts(negative) for a total of five volts across the capacitors C1 and/or C2.Thus, when one volt is applied at the node 606, the node 628 swings fourvolts consistent with the Miller effect. As a result, the currentflowing through the capacitors C1 and/or C2 is five times more than aconventional implementation because the voltage across the capacitor(s)is five times larger. Thus, more capacitance is seen from Vin withoutadding a large capacitor because more current flows through thecapacitor(s).

In one aspect, one or more notches are established in the feedback path(e.g., the first feedback path and/or the second feedback path) based onthe capacitor(s) (e.g., C1) to form a bandpass filter. At the higherfrequency, more capacitance is provided, which causes the frequencyresponse curve to be sharper and creates a sharper filter, asillustrated in FIG. 5B. The sharpening of the frequency response occursas a result of applying Miller effect to the second capacitor C2 whilecreating high-order filtering. The value of the capacitor C1 depends onR_(S), which is the series resistance of the Thévenin equivalent circuit602 and a desired bandwidth. For example, if the desired bandwidth isten (10) MHz and the series resistance R_(S) is one hundred (100) ohms,the capacitance C1 is eight (8) pF with Miller amplification of −4. Thevalue of the capacitor C2 depends on a high pass filter zero frequencyof the high pass filtering path, the series resistance R_(S) and thedesired bandwidth. For example, the capacitance of the capacitor C2 canbe fifteen (15) pico farad for a zero at 5.3 MHz.

In this aspect illustrated in FIG. 6, the high pass filtering pathincludes a filter with one zero. The high pass filtering path is used inconjunction with a lowpass filter to form a bandpass filter. A high passfilter feedback is provided to the first feedback path to create asharper frequency response. For example, the high pass filter feedbackis provided to the second capacitor C2, which is only visible at thehigher frequency.

FIG. 7 illustrates an N-path filter 700 including two high passfiltering paths according to aspects of the present disclosure. Forillustrative purposes, only a single branch of the N-path filter 700 isshown. The N-path filter 700, however, may include multiple branchesoperating in accordance with different phases (e.g., zero degrees,ninety degrees, one hundred and eighty degrees, two hundred and seventydegrees, etc.)

For illustrative purposes, some of the labelling and numbering of thedevices and features of FIG. 7 are similar to those of FIG. 6. Inaddition to the features of FIG. 6, FIG. 7 includes another high passfiltering path (or third feedback path). The third feedback pathincludes a third capacitor C3 in series with a second buffer and asecond high pass filter. The second buffer includes a fifth transistorM1 e and a sixth transistor M1 f. The second high pass filter includes asecond high pass capacitor CHP2 and a second resistor RHP2.

A terminal of the third capacitor C3 is coupled to a same node 622 as aterminal of the second capacitor C2. Another terminal of the thirdcapacitor C3 is coupled to a node 726, which is common to thetransistors M1 e and M1 f. For example, a source of the fifth transistorM1 e and a drain of the sixth transistor M1 f are coupled to each othervia the node 726. The high pass filter is coupled between the gate ofthe fifth transistor M1 e and the node 626. For example, a node 716between the second high pass capacitor CHP2 and the second resistor RHP2is also coupled to the gate of the fifth transistor M1 e.

Similar to the second capacitor C2, the third capacitor C3 is afrequency dependent capacitor. However, the third capacitor C3 is onlyvisible at a frequency that is higher than a frequency at which thesecond capacitor C2 is visible. The additional high pass filtering pathintroduces another zero (for a total of two zeros), which allows thepath through the third capacitor C3 to provide more out-of-bandrejection as illustrated in FIG. 8.

FIG. 8 is a graph 800 showing a first frequency response 802 of a lowerorder bandpass filter and a second frequency response 804 of a higherorder bandpass filter. For example, the lower order bandpass filter maycorrespond to the N-path filter 600 of FIG. 6 with a single zeroassociated with the high pass filtering path. The higher order bandpassfilter may correspond to the N-path filter 700 of FIG. 7 with two zerosassociated with the high pass filtering path. A radio frequencybandwidth 803 of both filters is the same. For example, both the lowerbandpass filter and the higher bandpass filter, in this case, have asame radio frequency bandwidth 803 (e.g., twenty megahertz (MHz)). Foran offset of forty megahertz (40 MHz), the higher order bandpass filterachieves four decibels (4 dB) more rejection than the lower orderbandpass filter.

FIG. 9 represents a Thévenin equivalent circuit 900 of the N-path filter700 including two high pass filtering paths according to aspects of thepresent disclosure. The Thévenin equivalent circuit 900 illustrates abaseband equivalent model of the N-path filter 700 as seen from anantenna represented by the Thévenin equivalent circuit 602. The firsthigh pass filtering path and the second high pass filtering path havetransfer functions in an S domain represented by H(s). An s-plane is acomplex plane on which Laplace transforms are graphed. It is amathematical domain where, instead of viewing processes in the timedomain modelled with time-based functions, they are viewed as equationsin the frequency domain. For example, the transfer function H(s) may berepresented by the following equation:

${H(s)} = \frac{0.8 \cdot S}{S + \frac{1}{\left( {R_{B}C_{B}} \right)}}$

where S is a complex frequency parameter for Laplace transforms, RBrepresents a high pass resistor (e.g., the first resistor RHP1 or thesecond resistor RHP2), and CB represents a high pass capacitor (e.g.,the first high pass capacitor CHP1 or the second high pass capacitorCHP2).

The gain of the N-path filter 700 may be calculated as follows:

$\frac{V_{1}(s)}{V_{in}(s)} = \frac{1 + {4 \cdot R_{SW} \cdot {s\left\lbrack {{\left( {1 + A_{V}} \right)C_{1}} + {\left( {1 + {A_{V}{H(s)}}} \right)C_{2}} + {\left( {1 + {A_{V}{H^{2}(s)}}} \right)C_{3}}} \right\rbrack}}}{\begin{matrix}{1 + {4 \cdot \left( {R_{S} + R_{SW}} \right) \cdot}} \\{s\left\lbrack {{\left( {1 + A_{V}} \right)C_{1}} + {\left( {1 + {A_{V}{H(s)}}} \right)C_{2}} + {\left( {1 + {A_{V}{H^{2}(s)}}} \right)C_{3}}} \right\rbrack}\end{matrix}}$

where V₁(s) is an output voltage of the N-path filter 700, R_(SW) is aswitch resistance of a branch switch (e.g., the branch resistance of thetransistors M1 a), R_(S) is the series resistance of the Théveninequivalent circuit 602, and A_(V) is a gain of the Miller amplifier. Itis to be noted that as the series resistance R_(S) becomes larger, thetransmit rejection improves.

FIG. 10 illustrates an N-path filter 1000 with multiple branches, witheach branch including two high pass filtering paths. For illustrativepurposes, some of the labelling and numbering of the devices andfeatures of FIG. 10 are similar to those of FIG. 7. However, instead ofa single branch as shown in FIG. 7, the N-path filter 1000 includesmultiple branches. For example, the multiple branches may include afirst branch 1010 and a second branch 1030. Each of the first branch1010 and the second branch 1030 are similar to the single branch shownin FIG. 7. However, for illustrative purposes, only a portion of thesecond branch 1030 is shown. For example, the second branch 1030 showstransistors M2 a and M2 b and the first capacitor C1. The first branch1010 may operate in accordance with a first phase (e.g., zero degrees)and the second branch 1030 may operate in accordance with a second phase(e.g., ninety degrees).

In this aspect, the first terminal of the first capacitor C1 is coupledto both the node 606 (associated with the input voltage) and the inputof the radio frequency Miller amplifier 620 via the first transistor orswitch Mla. A second terminal of the first capacitor C1 is coupled tothe output of the radio frequency Miller amplifier 620 via the secondtransistor M1 b. The first transistor M1 a and the second transistor M1b are configured as mixers to up-convert or down-convert a radiofrequency signal through the N-path filter 1000.

A local oscillator (not shown) is coupled to a gate of the firsttransistor M1 a and to a gate of the second transistor M1 b to configureeach of the transistors to operate as a mixer. The first branch 1010 andthe second branch 1030 are selectively coupled to a shared circuit node(e.g., node 606) that receives the input voltage Vin. In this aspect,however, the Miller amplifier 620 is selectively shared across the firstbranch 1010 and the second branch 1030. Thus, only one Miller amplifiersupports both branches.

FIG. 11 illustrates an N-path filter 1100 with multiple branches, witheach branch including two high pass filtering paths. For illustrativepurposes, some of the labelling and numbering of the devices andfeatures of FIG. 11 are similar to those of FIG. 7.

Instead of a single branch as shown in FIG. 7, the N-path filter 1100includes multiple branches. Additionally, FIG. 11 uses a baseband Milleramplifier for baseband amplification instead of the radio frequencyamplifier of FIG. 7 for radio frequency amplification. For example, themultiple branches may include a first branch 1110 and a second branch1130. Each of the first branch 1110 and the second branch 1130 issimilar to the single branch shown in FIG. 7. The first terminal of theMiller amplifier 620 is coupled or connected to the node 622 at a sourceof the transistor M1 a and the first terminal of the first capacitor C1of the first branch 1110. The second terminal of the Miller amplifier620 is coupled or connected to the node 624 at the second terminal ofthe first capacitor C1 of the first branch 1110. However, an out of bandsignal (e.g., a TX jammer) after the Miller amplifier 620, does not seea mixer (e.g., transistor M1 a).

For illustrative purposes, only a portion of the second branch 1130 isshown. For example, the second branch 1130 shows the transistor M2 a(without a transistor M2 b) and the first capacitor C1. Similar to thefirst branch 1110, a first terminal of a Miller amplifier 1120 of thesecond branch 1130 is coupled or connected to a node 1122 at a source ofthe transistor M2 a and a first terminal of the first capacitor C1 ofthe second branch 1130. A second terminal of the Miller amplifier 1120is coupled or connected to a node 1124 at a second terminal of the firstcapacitor C1 of the second branch 1130. The first branch 1110 mayoperate in accordance with a first phase (e.g., zero degrees) and thesecond branch 1130 may operate in accordance with a second phase (e.g.,ninety degrees).

The first terminal of the first capacitor C1 of the first branch 1110 iscoupled to the node 606 (associated with the input voltage Vin) via thefirst transistor or switch M1 a. A second terminal of the firstcapacitor C1 is coupled to the output of the radio frequency Milleramplifier 620.

The first transistor M1 a is configured as mixer. A local oscillator(not shown) is coupled to the gate of the first transistor M1 a toconfigure the transistor to operate as a mixer. The first branch 1110and the second branch 1130 are selectively coupled to a shared circuitnode (e.g., node 606) associated with the input voltage Vin. In thisaspect, however, the Miller amplifier 620 and the Miller amplifier 1120are respectively allocated to the first branch 1110 and the secondbranch 1130. Thus, a Miller amplifier is allocated for each of thebranches. Although not show, it is also contemplated that Milleramplifiers are provided as shown in FIG. 11 and also as shown in FIGS. 6and 7. In this case, the Miller amplifiers are selectively engaged,depending on which mode is selected.

FIG. 12 depicts a simplified flowchart of a method 1200 for filtering awireless signal at a receiver (e.g., of a user equipment). At block1202, a radio frequency signal is received at a shared circuit nodeselectively coupled to each of multiple branches of an N-path filter.Each branch of the N-path filter includes a first feedback pathincluding a Miller amplifier and a second feedback path having a node incommon with the first feedback path. At block 1204, a high impedance isgenerated at the common node to prevent the radio frequency signal fromtraversing the first feedback path and the second feedback path when theradio frequency signal is received at a first frequency. At block 1206,a low impedance at the common node is generated to allow the radiofrequency signal to traverse the first feedback path and/or the secondfeedback path when the radio frequency signal is received at a secondfrequency, the second frequency higher than the first frequency.

According to one aspect of the present disclosure, an N-path filter isdescribed. The filter includes first, second and third means forgenerating impedance. The first impedance generating means may, forexample, be the first capacitor C1. The second impedance generatingmeans may, for example, be the second capacitor C2. The third impedancegenerating means may, for example, be the third capacitor C3. In anotheraspect, the aforementioned means may be any module or any apparatus ormaterial configured to perform the functions recited by theaforementioned means.

FIG. 13 is a block diagram showing an exemplary wireless communicationsystem in which a configuration of the disclosure may be advantageouslyemployed. For purposes of illustration, FIG. 13 shows three remote units1320, 1330, and 1350 and two base stations 1340. It will be recognizedthat wireless communication systems may have many more remote units andbase stations. Remote units 1320, 1330, and 1350 include IC devices1325A, 1325B, and 1325C that include the disclosed N-path filter. Itwill be recognized that other devices may also include the disclosedN-path filter, such as the base stations, switching devices, and networkequipment. FIG. 13 shows forward link signals 1380 from the base station1340 to the remote units 1320, 1330, and 1350 and reverse link signals1390 from the remote units 1320, 1330, and 1350 to base station 1340.

In FIG. 13, remote unit 1320 is shown as a mobile telephone, remote unit1330 is shown as a portable computer, and remote unit 1350 is shown as afixed location remote unit in a wireless local loop system. For example,a remote unit may be a mobile phone, a hand-held personal communicationsystems (PCS) unit, a portable data unit such as a personal digitalassistant (PDA), a GPS enabled device, a navigation device, a set topbox, a music player, a video player, an entertainment unit, a fixedlocation data unit such as a meter reading equipment, or othercommunications device that stores or retrieves data or computerinstructions, or combinations thereof. Although FIG. 13 illustratesremote units according to the aspects of the disclosure, the disclosureis not limited to these exemplary illustrated units. Aspects of thedisclosure may be suitably employed in many devices, which include theN-path filter.

For a firmware and/or software implementation, the methodologies may beimplemented with modules (e.g., procedures, functions, and so on) thatperform the functions described herein. A machine-readable mediumtangibly embodying instructions may be used in implementing themethodologies described herein. For example, software codes may bestored in a memory and executed by a processor unit. Memory may beimplemented within the processor unit or external to the processor unit.As used herein, the term “memory” refers to types of long term, shortterm, volatile, nonvolatile, or other memory and is not to be limited toa particular type of memory or number of memories, or type of media uponwhich memory is stored.

If implemented in firmware and/or software, the functions may be storedas one or more instructions or code on a computer-readable medium.Examples include computer-readable media encoded with a data structureand computer-readable media encoded with a computer program.Computer-readable media includes physical computer storage media. Astorage medium may be an available medium that can be accessed by acomputer. By way of example, and not limitation, such computer-readablemedia can include RAM, ROM, EEPROM, CD-ROM or other optical diskstorage, magnetic disk storage or other magnetic storage devices, orother medium that can be used to store desired program code in the formof instructions or data structures and that can be accessed by acomputer; disk and disc, as used herein, includes compact disc (CD),laser disc, optical disc, digital versatile disc (DVD), floppy disk andBlu-ray disc where disks usually reproduce data magnetically, whilediscs reproduce data optically with lasers. Combinations of the aboveshould also be included within the scope of computer-readable media.

In addition to storage on computer-readable medium, instructions and/ordata may be provided as signals on transmission media included in acommunication apparatus. For example, a communication apparatus mayinclude a transceiver having signals indicative of instructions anddata. The instructions and data are configured to cause one or moreprocessors to implement the functions outlined in the claims.

The various illustrative logical blocks, modules, and circuits describedin connection with the disclosure herein may be implemented or performedwith a general-purpose processor, a digital signal processor (DSP), anapplication specific integrated circuit (ASIC), a field programmablegate array (FPGA) or other programmable logic device, discrete gate ortransistor logic, discrete hardware components, or any combinationthereof designed to perform the functions described herein. Ageneral-purpose processor may be a microprocessor, but in thealternative, the processor may be any conventional processor,controller, microcontroller, or state machine. A processor may also beimplemented as a combination of computing devices, e.g., a combinationof a DSP and a microprocessor, multiple microprocessors, one or moremicroprocessors in conjunction with a DSP core, or any other suchconfiguration.

Although the present disclosure and its advantages have been describedin detail, it should be understood that various changes, substitutions,and alterations can be made herein without departing from the technologyof the disclosure as defined by the appended claims. For example,relational terms, such as “above” and “below” are used with respect to asubstrate or electronic device. Of course, if the substrate orelectronic device is inverted, above becomes below, and vice versa.Additionally, if oriented sideways, above and below may refer to sidesof a substrate or electronic device. Moreover, the scope of the presentapplication is not intended to be limited to the particularconfigurations of the process, machine, manufacture, and composition ofmatter, means, methods, and steps described in the specification. As oneof ordinary skill in the art will readily appreciate from thedisclosure, processes, machines, manufacture, compositions of matter,means, methods, or steps, presently existing or later to be developedthat perform substantially the same function or achieve substantiallythe same result as the corresponding configurations described herein maybe utilized according to the present disclosure. Accordingly, theappended claims are intended to include within their scope suchprocesses, machines, manufacture, compositions of matter, means,methods, or steps.

What is claimed is:
 1. A N-path filter, comprising: a first branchcomprising: a first feedback path comprising: an amplifier having aninput coupled to an input voltage, and a first capacitor coupled to boththe input voltage and an output of the amplifier; and a second feedbackpath having a node in common with the first feedback path, the secondfeedback path comprising: a first high pass filter coupled to the outputof the amplifier and a second capacitor coupled to both the firstcapacitor and the first high pass filter.
 2. The N-path filter of claim1, further comprising a buffer between the first high pass filter andthe second capacitor.
 3. The N-path filter of claim 2, in which thebuffer comprises a biased source follower device.
 4. The N-path filterof claim 1, further comprising: a third feedback path merged with thesecond feedback path, the third feedback path comprising: a second highpass filter coupled to an output of the first high pass filter, and athird capacitor coupled to both the first capacitor and the secondcapacitor and to the second high pass filter.
 5. The N-path filter ofclaim 4, further comprising a buffer between the first high pass filterand the second capacitor, the buffer coupled between an output of thefirst high pass filter and an input of the second high pass filter. 6.The N-path filter of claim 1, integrated into a receive (RX) chainbefore or after a low noise amplifier (LNA) in accordance with a shuntconfiguration.
 7. The N-path filter of claim 1, in which the amplifiercomprises a Miller amplifier or an amplifier based at least in part onMiller effect.
 8. The N-path filter of claim 7, in which the Milleramplifier comprises a radio frequency Miller amplifier and in which, afirst terminal of the first capacitor is coupled to both the inputvoltage and the input of the Miller amplifier via a first switch; and asecond terminal of the first capacitor is coupled to the output of theMiller amplifier via a second switch.
 9. The N-path filter of claim 7,in which the Miller amplifier comprises a baseband Miller amplifier andin which, a first terminal of the first capacitor is coupled to a firstterminal of a first switch and the input of the Miller amplifier; and asecond terminal of the first capacitor is coupled to the output of theMiller amplifier, a second terminal of the first switch coupled to theinput voltage.
 10. The N-path filter of claim 1, further comprising, asecond branch, in which the first branch and the second branch areselectively coupled to a shared circuit node that receives the inputvoltage, in which the first branch and the second branch arerespectively operated in accordance with a first phase and a differentsecond phase, and in which the amplifier is selectively shared acrossthe first branch and the second branch.
 11. The N-path filter of claim1, further comprising a second branch, in which the first branch and thesecond branch are selectively coupled to a shared circuit node thatreceives the input voltage, in which the first branch and the secondbranch are respectively operated in accordance with a first phase and adifferent second phase, and in which the amplifier is allocated to thefirst branch and another amplifier is allocated to the second branch.12. A method for filtering a wireless signal at a receiver comprising:receiving a radio frequency signal at a shared circuit node selectivelycoupled to each of a plurality of branches of an N-path filter, in whicheach branch of the N-path filter comprises a first feedback pathincluding an amplifier and a second feedback path having a common nodewith the first feedback path; generating a high impedance at the commonnode to prevent the radio frequency signal from traversing the firstfeedback path and/or the second feedback path when the radio frequencysignal is received at a first frequency; and generating a low impedanceat the common node to allow the radio frequency signal to traverse thefirst feedback path and/or the second feedback path when the radiofrequency signal is received at a second frequency higher than the firstfrequency.
 13. The method of claim 12, in which the first frequencycomprises an in-band frequency that is close to a direct current (DC)frequency and the second frequency comprises an out-of-band frequency.14. The method of claim 12, further comprising down-converting the radiofrequency signal of the first frequency.
 15. The method of claim 12,further comprising: amplifying the radio frequency signal received atthe second frequency using the amplifier; down-converting the radiofrequency signal that is amplified; providing the radio frequency signalthat is down-converted to an up-converter through the low impedancegenerated by a first capacitor and a second capacitor associated withthe first feedback path and the second feedback path; and up-convertingthe radio frequency signal and providing the radio frequency signal thatis up-converted to the shared circuit node.
 16. The method of claim 12,in which the amplifier comprises a Miller amplifier or an amplifierbased at least in part on Mille effect.
 17. A N-path filter, comprising:a first branch comprising: a first feedback path comprising: anamplifier having an input coupled to an input voltage, and first meansfor generating an impedance, the first impedance generating meanscoupled to both the input voltage and an output of the amplifier; and asecond feedback path having a node in common with the first feedbackpath, the second feedback path comprising: a first high pass filtercoupled to the output of the amplifier and second means for generatingan impedance, the second impedance generating means coupled to both thefirst impedance generating means and the first high pass filter.
 18. TheN-path filter of claim 17, further comprising a buffer between the firsthigh pass filter and the second impedance generating means.
 19. TheN-path filter of claim 18, in which the buffer comprises a biased sourcefollower device.
 20. The N-path filter of claim 17, further comprising:a third feedback path merged with the second feedback path, the thirdfeedback path comprising: a second high pass filter coupled to an outputof the first high pass filter, and third means for generating impedance,the third impedance generating means coupled to both the first impedancegenerating means and the second impedance generating means and to thesecond high pass filter.
 21. The N-path filter of claim 20, furthercomprising a buffer between the first high pass filter and the secondimpedance generating means, the buffer coupled between an output of thefirst high pass filter and an input of the second high pass filter. 22.The N-path filter of claim 17, integrated into a receive (RX) chainbefore or after a low noise amplifier (LNA) in accordance with a shuntconfiguration.
 23. The N-path filter of claim 17, in which the amplifiercomprises a Miller amplifier or an amplifier based at least in part onMille effect.